High Gain Detector Techniques for Low Bandwidth Low Noise Phase-Locked Loops

ABSTRACT

In described examples, a feedback loop has phase detection (PD) circuitry that has a reference input to receive a reference frequency signal, a feedback input to receive a feedback signal, and phase difference outputs. A phase to digital converter (P2DC) includes a first phase to charge converter (PCC) that has a gain polarity and a first phase error output; a second PCC that has an opposite gain polarity and a second phase error output. A differential loop filter has an amplifier with an inverting input coupled to the first phase error output and a non-inverting input coupled to the second phase error output. An analog to digital converter (ADC) has an input coupled to an output of the differential loop filter. A feedback path is coupled to the output of the P2DC, with an output of the feedback path providing the feedback signal to the PD feedback input.

CROSS-REFERENCE TO RELATED APPLICATION

This application claims priority to U.S. Provisional Patent ApplicationNo. 63/136,245 filed Jan. 12, 2021, the entirety of which isincorporated herein by reference.

TECHNICAL FIELD

This relates to high gain phase detector techniques for a low noisefeedback loop.

BACKGROUND

Low phase noise operation for phase-locked loops (PLLs) or relatedfeedback structures is enabled by high gain phase detector (PD)techniques. A high gain PD allows low detector noise to be achieved,which is typically a key bottleneck to achieving low phase noise at lowfrequency offsets.

There are several techniques for achieving high gain PD functionality.An example is a slope-based sampling PD structure, see, for example: “A28-nm 75-fsrms Analog Fractional-N Sampling PLL With a Highly Linear DTCIncorporating Background DTC Gain Calibration and Reference Clock DutyCycle Correction,” Wanghua Wu et al, 2019. Another example is an Up/Downresistor-capacitor (RC) charging circuit that utilize a limited timerange for the Up/Dn timing window, see, for example: “A Low Area,Switched-Resistor Based Fractional-N Synthesizer Applied to a MEMS-BasedProgrammable Oscillator Phase detector,” Michael H. Perrott, et al,2010. The slope-based sampling PD structure offers high gain but suffersfrom process and temperature (PT) sensitivity of that gain since theslope will generally be impacted by PT variations. The Up/Dn RC chargingcircuits offer gain that is generally robust against PT variation butare generally more limited by supply voltage than the slope-basedstructure. Both approaches are sensitive to supply noise.

SUMMARY

In described examples, a feedback loop includes a fully differentialloop filter structure with gain fed by high gain phase detectors withopposite gain polarity for supply rejection. In some examples, adifferential output is fed into differential or pseudo-differential ADC.In some examples, the feedback loop is augmented with bang-bang detectorwith a dead zone in order to extend phase error range for faster phaselocking.

In described examples for low bandwidth applications, an enhanceddelta-sigma modulator is used to reduce quantization noise at lowfrequencies without significant impact to shaped noise at highfrequencies

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 is a block diagram of an example phase locked loop (PLL).

FIG. 2 is an example noise model of the PLL of FIG. 1.

FIGS. 3-5 are plots of phase noise level (dBc/Hz) versus offsetfrequency (f) for the noise model of FIG. 2.

FIG. 6A is a schematic and FIG. 6B is a timing diagram of a prior artphase detector and loop filter.

FIG. 7A is a schematic and FIG. 7B is a timing diagram of another priorart phase detector.

FIG. 8A is a schematic and FIG. 8B is a timing diagram of another priorart phase detector.

FIG. 9A is a schematic and FIG. 9B is a timing diagram of an examplehigh gain phase detector.

FIG. 10 is a schematic of an equivalent circuit for the phase detectorof FIG. 9A.

FIG. 11A is a schematic and FIG. 11B is a timing diagram of an examplephase to charge converter without charge pump boosting.

FIG. 12 is a timing diagram showing the effect of charge pump structuresprovided in the phase detector of FIG. 9A.

FIG. 13A is a schematic and FIG. 13B is a timing diagram of anotherexample high gain phase detector.

FIG. 14 is a schematic diagram of an example differential high gainphase detector and loop filter.

FIG. 15 is a schematic of a simple XOR phase detector.

FIG. 16 is a schematic diagram of an example XOR differential high gainphase detector and loop filter.

FIG. 17 is a schematic of an example phase detector and frequencydetector circuit.

FIG. 18 is a timing diagram illustrating operation of the examplecircuit of FIG. 17.

FIG. 19 is a plot illustrating phase noise in an example PLL.

FIG. 20 is a block diagram of an example voltage supply for a PLL.

FIG. 21 is a plot of noise vs frequency for the supply of FIG. 20.

FIG. 22 is a block diagram of an example feedback loop with high gainphase detector.

FIG. 23 is a timing diagram illustrating operation of the high gainphase detector within the feedback loop of FIG. 22.

FIG. 24 is a schematic of an example phase to digital converter for alow BW feedback loop.

FIG. 25 is a simulation model for the resistor switching section of FIG.24.

FIG. 26 is a schematic of an example fully differential phase to digitalconverter for a low BW feedback loop.

FIG. 27 is a schematic of another example fully differential phase todigital converter for a low BW feedback loop.

FIG. 28 is a schematic of an example alternative switched resistor phaseto charge converter.

FIG. 29 is a schematic of an example alternative switch scheme.

FIG. 30 is a schematic illustrating example configurability options foran example phase to digital converter.

FIG. 31 is a block diagram and FIG. 32 is a timing diagram of an examplecircuit to generate early/late pulses.

FIGS. 33-36 are schematics and timing diagrams for example linear phasedetectors.

FIGS. 37A-37E are timing diagrams illustrating example bang-bang timingwith linear phase detector timing.

FIG. 38 is a schematic of an example circuit to generate bang-bangsignals.

FIG. 39 is a block diagram of an example 2^(nd) order MASH delta-sigmamodulator.

FIG. 40 is a block diagram of an example enhanced 2^(nd) order MASHdelta-sigma modulator.

FIG. 41 is an example noise model of the example enhanced delta-sigma ofFIG. 40.

FIG. 42 is a plot illustrating simulation results for the enhanceddelta-sigma of FIG. 40.

FIG. 43 is a block diagram of an example analog phase locked loopcontrolled by the feedback loop of FIG. 22.

FIG. 44 is a block diagram of the example analog phase locked loop ofFIG. 43 augmented by a digital PLL.

FIG. 45 is a plot illustrating phase noise level (dBc/Hz) versus offsetfrequency for simulated operation of example noise model of FIG. 2.

FIG. 46 is a plot illustrating phase noise level (dBc/Hz) versus offsetfrequency for simulated operation of example system 4300 of FIG. 43.

FIGS. 47A, 47B are plots illustrating operation of an example bang-bangcircuit.

DETAILED DESCRIPTION

In the drawings, like elements are denoted by like reference numeralsfor consistency.

In examples described herein, achievement of low phase noise at lowoffset frequencies for phase-locked loops (PLLs) or related feedbackstructures is enabled by high gain phase detector (PD) techniques. Inparticular, a high gain PD allows reduction of the impact of detectornoise to be achieved, which is typically a key bottleneck to achievinglow phase noise at low frequency offsets.

In other examples described herein, low noise PLLs or related feedbackstructures are greatly aided by achieving a wide bandwidth (BW) for thePLL in order to suppress voltage-controlled oscillator (VCO) noise.However, wide BW PLLs are significantly impacted by phase detectornoise, and therefore must achieve low phase detector noise in order toachieve low jitter. High gain phase detector techniques allow lowdetector noise impact to be achieved.

While high gain PD techniques exist, they are generally sensitive toprocess and temperature (PT) variation, voltage supply noise, and/orlimited supply voltage (often <1.2V for core devices in advanced CMOS).PD gain variations can degrade PLL jitter performance across PTvariation due to corresponding changes in the PLL bandwidth.

Voltage supply noise can degrade the low frequency phase noiseperformance. While such supply noise can be reduced with passive lowpassfiltering, such filters require substantial area and may even requireinclusion of undesired off-chip components such as discrete capacitors.Lower supply voltage is desired to reduce power consumption and allowuse of core devices in advanced CMOS but can degrade PLL performance dueto reduced PD gain.

There are several techniques for achieving high gain PD functionality.An example is a slope-based sampling PD structure. Another example is anUp/Down resistor-capacitor (RC) charging circuit that utilizes a limitedtime range for the Up/Dn timing window. The slope-based sampling PDstructure offers high gain but suffers from PT sensitivity of that gainsince the slope will generally be impacted by PT variations. The Up/DnRC charging circuits offer gain that is generally robust against PTvariation but are generally more limited by supply voltage than theslope-based structure. Both approaches are sensitive to supply noise.

In examples described herein, high gain PD techniques reduce sensitivityto supply noise by leveraging a differential structure. In anotherexample, a technique is described for augmenting a delta-sigma modulatorto reduce its low frequency quantization noise without substantiallyincreasing high frequency quantization noise, which is useful forimproving low frequency phase noise performance without incurringadditional noise folding due to nonlinearity of the phase detector. Inanother example, a digital-to-time converter is used as an alternativefor reducing the quantization noise with the benefit of enabling widerbandwidth, but comes at the cost of higher complexity, power, and area.

Examples described herein are based on improvements to the Up/Dn RCcharging circuit approach to achieve higher PD gain and to reducesensitivity to supply noise. A higher PD gain is achieved by leveragingcharge pump techniques to increase the effective supply voltage seen bythe PD during the Up/Dn charge/discharge times.

A lower sensitivity to supply noise is achieved through loop filtertopologies that may be combined with various phase detector techniques.Single-ended and differential versions of example loop filter topologiesare described herein. Some examples described herein utilize adifferential structure in order to reduce sensitivity to supply noisewhile maintaining high gain for the PD. In some described examples, anADC is included to digitize the differential signal.

In general, achievement of lower noise through brute force methods suchas increased power/area encounter practical limits due power/areaconstraints for a product. In examples described herein, techniques toincrease PD gain utilize circuit topologies that can be implemented withmodest power/area requirements and enable state-of-the art jitterrequirements to be met. Achievement of insensitivity to low frequencysupply noise can often be achieved with external capacitors, but this isundesirable due to increased cost to the final system and difficultiesin board design to avoid noise injection into the routing traces andpins associated with the external capacitors. Examples described hereinuse techniques for reducing supply sensitivity that avoid the need forsuch external capacitors.

FIG. 1 is a block diagram of an example phase locked loop (PLL) 100. Avoltage-controlled oscillator (VCO) 108 outputs a variable frequencysignal on oscillator output node 122 that is tuned according to acontrol voltage 107. Feedback is used to lock the VCO output frequencyto a multiple of the reference frequency input signal 120 through theuse of a multi-modulus frequency divider (MMD) 110, phase detector (PD)102, and loop filter 106. In this example, the phase detector 102 alsoincludes frequency detection (FD) logic 104. Phase detector 102 includespulse generation (PG) logic 103 that produces up and down pulses whosepulse width varies with the phase difference between the referencefrequency 120, Ref, and divider (Div) output 121. A phase to chargeconverter (PCC) 105 converts the up and down PD signals into pulses thatare then filtered by loop filter 106 to form the control voltage 107.PCC 105 is configured to provide a high gain for PD 102

Digital-to-time converter (DTC) 112 is utilized to reduce quantizationerror from delta-sigma modulator 114 dithering of divider 110 so as toavoid noise folding due to nonlinearity of the high gain PD 102. In someexamples, DTC 112 allows a phase adjustment. DTC 112 produces a variabledelay that is determined by a digital input value provided by deltasigma 114 and MMD 110. The frequency divide value of MMD 112 iscontrolled by delta sigma 114. The output of MMD 110 serves as a clockinput to delta sigma 114 and an input to DTC 112.

In some examples, the divisor value of multi-modulus divider 110 may bechanged dynamically.

FIG. 2 is an example noise model 200 of the PLL of FIG. 1. FIG. 3 is aplot of raw phase noise level (dBc/Hz) versus offset frequency (f) forthe noise model of FIG. 2. FIG. 3 is a plot of phase noise level beforefiltering within PLL 100 (FIG. 1). FIG. 4 is a plot of phase noise levelafter filtering within PLL 100 (FIG. 1). This example model includesphase detector 202, loop filter 206, VCO 208, and divider 210. K_(D) 202is the gain of phase detector 102. H(s) 206 is the transfer function ofloop filter 106. (2πKv)/s 208 is the transfer function for VCO 108. N isthe divider value of divider 110.

Various sources of noise contribute to degradation of loop performance,such as: phase detector noise 221, quantization noise from the divider,DTC thermal noise and delta sigma dithering noise 223, some residualnoise that is not canceled, supply noise, etc. Supply noise affects allblocks, but especially is an issue for the phase detector and loopfilter. Delta sigma modulator 114 causes noise 223 to rise to higherfrequencies that can be filtered by the loop filter 206. VCO noise 222gets high-pass filtered by the loop, but some low frequency noise getsthrough. Phase detector 202 is low pass filtered by the loop filter 206but some high frequency noise gets through.

As illustrated in FIG. 4, after filtering by PLL 100, detector noise 421dominates at low frequency offsets relative to the PLL bandwidthindicated at 301. VCO noise 422 dominates at high frequency offsets. DTCand DS noise 422 is reduced due to filtering by the PLL.

Referring to FIG. 2, expression (1) quantifies the transfer functionrelationship from detector noise to the output signal on output terminal122. For the case where s is much less than PLL bandwidth (BW),expression (1) can be simplified to expression (2).

$\begin{matrix}{{H_{{out}/\det}(s)} = \frac{{H(s)}2\;\pi\;{{Kv}/s}}{1 + {{K_{D}/{{NH}(s)}}\; 2\pi\;{{Kv}/s}}}} & (1) \\{\mspace{104mu}{\approx {\frac{N}{K_{D}}\mspace{14mu}{for}\mspace{14mu} s} ⪡ {{PLL}\mspace{14mu}{BW}}}} & (2)\end{matrix}$

Thus, detector noise 421 is approximately equal to N/K_(D), therefore,maximizing detector gain K_(D) results in minimizing the impact ofdetector noise on output 122.

FIG. 5 illustrates noise folding that may occur if DTC 112 (see FIG. 1)is not included in PLL 100. Without DTC 112, nonlinearity in phasedetector 102 leads to noise folding of delta sigma noise 423, asindicated at 523. Such noise folding is avoided by the use of DTC 112which reduces the impact of dithering by delta sigma modulator 114 (seeFIG. 1) on phase error.

FIG. 6A is a schematic of a prior art phase detector 601 and loop filter602 for use in a feedback structure such as a phase locked loop. Phasedetector 601 generates an up-pulse signal 624 and a down-pulse signal625 whose widths are a function of phase difference between referencefrequency signal 620 and feedback divided signal 621, as illustrated inFIG. 6B.

Charge pump 626 is turned on in response to up pulse signal 624 andcharge pump 627 is turned on in response to down pulse signal 625.Charge pumps 626, 627 are added to allow up or down control of the VCOtuning voltage formed on node 628. Expression (3) represents the loopfilter transfer function, H(s), from output of the charge pump to theVCO tuning voltage 628. In general, larger charge pump current, which isadvantageous for improved detector noise, must be accompanied by anincrease in loop filter capacitors to achieve a given PLL bandwidth.This often leads to the requirement of large physical capacitors thattypically must be located off chip, which is undesirable for anintegrated single chip solution.

$\begin{matrix}{{H(s)} = {\frac{1}{s\left( {C_{1} + C_{2}} \right)}\frac{1 + {s/w_{z}}}{1 + {s/w_{p}}}}} & (3)\end{matrix}$

FIG. 7A is a schematic of a prior art phase detector 701 and loop filterfor use in a feedback structure such as a phase locked loop that doesnot use current sources to increase phase detector gain. Instead, theratio of the reference signal period (Tref) to the divided feedbacksignal period (Tdiv) is increased. In this example, the divisor isselected so that the feedback divided signal 721 frequency is four timesthe frequency of the reference signal 720. Phase detector 701 sees aphase error range over a smaller phase window and the correspondingphase detector gain is increased due to the small phase error range. Inthis example, there is a 4× improvement due to 4× frequency ratio. Thisprovides a relatively stable phase detector gain since the Tref/Tdivratio is PVT insensitive. However, this phase detector and loop filterapproach is very sensitive to supply voltage (Vdd) noise and there issome impact of noise folding for fractional-N implementations due tononlinearity from RC charging behavior within the loop filter.

Phase detector 701 generates an up-pulse signal 724 and a down-pulsesignal 725 whose widths are a function of phase difference betweenreference frequency signal 720 and feedback divided signal 721, asillustrated in FIG. 7B.

Phase detector 701 is based on an RC charging mechanism with resistor R1and capacitor C1. While up signal 724 is active capacitor C1 is chargedvia resistor R1. While down signal 725 is active C1 is discharged. Whenup or down are not present, then capacitor C1 holds the voltage.

FIG. 8A is a schematic and FIG. 8B is a timing diagram of another priorart switched resistor phase detector 801 and loop filter 802. In thisexample, only the phase to charge module 805 of the phase detector isillustrated in detail for simplicity. Pulse generation module 803generates the pulse signals illustrated in FIG. 8B. In this example, aseparate up-charge resistor Rup and down-charge resistor Rdn areconnected to charging node 828 and charging capacitor Cdet. In thisexample, the phase detector includes a pulse generation circuit 803 togenerate the up-pulse signal 824, down-pulse signal 825 and gate signal826.

In this example, the divider is configured to provide a divided feedbacksignal 821 that has a frequency that is lower than the frequency of thereference signal 820. As in the example of FIG. 7A, gain of the phasedetector is increased due to the ratio of Fref/Fdiv.

Switch 830 is controlled by gate signal 826 to only transfer charge fromRC node 828 to loop filter 802 for a limited period of time. Theresulting increase in phase detector gain reduces the impact of noisethat is transferred from phase to charge converter 805 to loop filter802.

Phase detectors 701 (FIG. 7A) and 802 are described in more detail in “ALow Area, Switched-Resistor Based Fractional-N Synthesizer Applied to aMEMS-Based Programmable Oscillator Phase detector,” Michael H. Perrott,et al, 2010.

High Bandwidth, High Gain Phase Detector Examples

FIG. 9A is a schematic of a switched resistor phase detector 901 andloop filter 902. In this example, the gain of phase detector 901 isimproved with a structure that alters RC charging. Phase detector gainis limited by the supply voltage. If a larger supply voltage is used,the gain can be increased. However, process limitations limit themagnitude of the supply voltage without device problems due toovervoltage issues. In this example, the “effective” supply voltage onthe RC charging circuit is raised by using a voltage boosting structure926, 927 (also referred to as a “charge pump”) that augments the phasedetector. To do this, capacitors Cup2 and Cdn2 are added along withinverters 926 and 927 in order to inject additional charge intocapacitors Cup1 and Cdn1 during assertion of Up and Dn signals,respectively. This injection of extra charge has a similar effect on RCcharging as what would be achieved with a higher supply voltage. Anextra switch 932 allows the Up and Dn network charge states to be sharedafter the Up and Dn charging events occur. Switches 933 and 934 pass thecombined Up and Dn charge states to capacitor C1 such that a phase errorvoltage signal (Vfilt) is achieved which can be further filtered beforeinfluencing the VCO control voltage Vctrl.

In described examples, charge pumps 926, 927 boost a charging voltage onboost capacitors Cup2, Cdn2, respectively in order to increase gain ofthe phase detector. In another example, a charge pump structure thatboosts a current into a suitable element, such as an inductor, may beused to increase an effective supply voltage to increase the gain of aphase detector.

FIG. 9B is a timing diagram illustrating timing signals 924, 925, 926generated by pulse generator module 903 in response to a referencesignal 920 (Ref_xN) and a feedback signal 921 (Div). The time durationof Up pulse 924 is proportional to the time between an edge of referencesignal 920 and feedback signal 921. The time duration of Dn pulse 925 isproportional to the time between an edge of reference signal 920 andfeedback signal 921. The total length of Up pulse 924 and Dn pulse 925is constrained to be the period of the reference signal, Tspan.

For example, when down-pulse 925 becomes asserted and switch 931 isclosed, the output of inverter 927 will transition to a high voltagestate and thereby charge capacitor Cdn2 through resistor Rdn to groundvia Vdn RC node 937 and also share charge with capacitor Cdn1. Then,when up-pulse 924 is asserted and switch 930 is closed, the output ofinverter 926 will become low and thereby charge capacitor Cup2 throughresistor Rup from Vreg via Vup RC node 936 and also share charge withcapacitor Cup1. Then, when up-pulse 924 and down-pulse 925 arede-asserted and switches 930, 931 are open, gate pulse 926 is activatedto close switches 932, 933 and 934 and thereby couple the RC nodes 936,937 to filter capacitor C1 at phase error output node 928

In this example, only the phase to charge converter 905 of the phasedetector 901 is illustrated in detail for simplicity. Pulse generationmodule 903 generates the pulse signals illustrated in FIG. 9B. Pulsegeneration circuit 903 generates the up-pulse signal 924, down-pulsesignal 925 and gate signal 926. In this example, down-pulse signal 925is enabled only from the rising edge of ref signal 920 to the risingedge of divide signal 921, and the up-pulse signal 924 is enabled onlyfrom the rising edge of divide signal 921 to the rising edge of refsignal 920. In this manner, up-pulse signal 924 and down pulse signal925 are non-overlapping and have a total active time that is equivalentthe period (Tspan) of ref signal 920.

In this example, the divider is configured to provide a divided feedbacksignal 921 that has a frequency that is lower than the frequency of thereference signal 920. Gain of the phase detector is increased due to theratio of Fref/Fdiv.

Switches 933, 934 are controlled by gate signal 926 to transfer chargefrom RC nodes 936, 937 to output node 928 for a limited period whilegate signal 926 is active. This prevents the phase error voltage Vfiltas well as the VCO control voltage Vctrl from being disturbed by the RCcharging activity during enablement of Up and Dn pulses.

Up-pulse 924 and down-pulse 925 are enabled while the gate switches 932,933, 934 are off. Phase detector gain is improved by an alpha factor,which is a ratio of the caps as given by expression (4). equation.Expression (5) represents the total gain factor of phase detector 901assuming capacitors Cup1 and Cdn1 are equal in value and that capacitorsCup2 and Cdn2 are also equal in value.

$\begin{matrix}{{\alpha_{\det^{=}}\frac{C_{{{up}\; 1}\;} + C_{{up}\; 2}}{C_{{up}\; 1}}} = \frac{C_{{dn}\; 1} + C_{{dn}\; 2}}{C_{{dn}\; 1}}} & (4) \\{\alpha_{\det}\frac{F_{Ref\_ xN}}{\pi\mspace{11mu} F_{Div}}\frac{V_{reg}}{2}} & (5)\end{matrix}$

In some examples, multiple capacitors may be provided that may beselectively switched off using switches, a multiplexor, or other knownor later developed technique to dynamically change the gain of thesystem by varying the capacitor ratio alpha to optimize the gain. If oneconsiders only minimization of the impact of detector noise, alphashould be selected to be as high as possible. However, otherconsiderations such as implementation area, achievable switching speedof inverters 926 and 927 with capacitive loading, and impact on supplymay impact the optimal setting of alpha.

FIG. 10 is a schematic of an equivalent circuit 1005 for the phase tocharge converter 905 for the phase detector 901 of FIG. 9A. As describedfor FIG. 9A, charge pump structures 926, 927 and Cup2, Cdn2 produce aneffect equivalent to raising the supply voltage. In this example, theresult is the same as if the supply voltage is raised by an amount equalto α_(det) times one half the supply voltage and the ground voltage islowered by an amount equal to α_(det) times one half the supply voltage.In this example, the supply voltage for phase detector 901 is aregulated voltage Vreg. In other examples, the supply voltage for thephase detector may be the chip wide supply voltage Vdd or anotherdifferent voltage source. In this example, the use of charge pumpstructures 926, 927 and Cup2, Cdn2 allows an effective increase in phasedetector gain as determined by the capacitor ratio α_(det).

FIG. 11A is a schematic and FIG. 11B is a timing diagram of an examplephase to charge converter (PCC) 1105 that is similar to PCC 905 (FIG.9A) without charge pump structures 926, 927. In this example withoutgain boosting charge pumps, discharge at Vup node 1136 during up-pulse924 and discharge at Vdn node 1137 during down-pulse 925 starts atapproximately Vreg/2 for both the up and down path. A resulting Vfiltsignal is formed on output node 1128.

FIG. 12 is a timing diagram of example PCC 905 showing the effect ofcharge pump structures 926, 927 and Cup2, Cdn2. In this example, gainboosting is achieved by increasing the initial voltage across the Rup,Rdn resistors during the enable times of up-pulse 924 and down-pulse 925as indicated on Vup 936 and Vdn 937 at 1242, 1243 respectively. Thisresults in a larger change in the amplitude of Vfilt 928 for a givenchange in phase error. Thus, PCC 905 with charge pump structures 926,927 and Cup2, Cdn2 has a higher gain than PCC 1105 without charge pumpstructures.

FIG. 13A is a schematic of another example high gain phase to chargeconverter (PCC) 1305 that is similar to PCC 905 (FIG. 9A). In thisexample, gate 932 is controlled by gate0 signal 13260, while gates 933,934 are controlled by an offset gate1 signal 13261.

FIG. 13B is a timing diagram illustrating timing signals 924, 925,13260, 13261 generated by a pulse generator module similar to PG 903(FIG. 9A) in response to a reference signal 920 (Ref_xN) and a feedbacksignal 921 (Div). Gate0 signal 13260 is offset from gate1 signal 13261by a small amount so that gate 932 is closed slightly before gates 933,934. This allows ripple on nodes 1336, 1337 to settle out prior toclosing switches 933, 934 and thereby reduces ripple sent to filter 1302and output Vctrl. In this example, the offset time is approximately5%-10% of Tspan, though optimal offset time will vary according tosettling behavior when switch 13260 is on as well as other constraints.Note that Gate0 signal is shown to become de-asserted before Gate1signal, but other implementations could have Gate0 signal becomede-asserted at the same time or after Gate1 signal.

FIG. 14 is a schematic diagram of an example differential high gainphase detector and loop filter 1400 that can be used in the example PLL100 of FIG. 1. In this example, a wide band feed-forward (FF) path 1401includes a high gain PCC cell 1410 coupled to FF filter 1412. Lossyintegrating path 1402 includes an opamp 1420 with an inverting input1421 coupled to receive a filtered output from PCC cell 1425 and anon-inverting input 1422 coupled to receive a filtered output from PCCcell 1426.

Lossy integrating path 1402 also includes a frequency detection path1424 in which switch 1411 is configured to couple inverting input 1421to ground through resistor Rfd_lo when a signal FDlo asserts in caseswhere the frequency of an output signal, such as Out signal 122 (FIG. 1)is too low and in which switch 1412 is configured to couple invertinginput 1421 to Vreg through resistor Rfd_hi when a signal FDhi asserts incases where the frequency of the output signal is too high.

FF filter 1412 combines the output 1407 of PCC cell 1410 and integratingpath 1402 to produce a control signal Vctrl on output node 1414. Controlsignal Vctrl is used to control a variable frequency oscillator thatproduces Out signal 122 (FIG. 1). In another example, additionalfiltering may be provided for control signal Vctrl before being outputon node 1414.

PCC cells 1410, 1425, and 1426 can be the same as PCC 905 (FIG. 9A), PCC1005 (FIG. 10), PCC 1105 (FIG. 11B), PCC 1305 (FIG. 13A) or anotherknown or later developed PCC cell. However, notice that the Up signal924 and Dn signal 925 are opposite between PCC 1425 and PVV 1426. Thisallows cancellation of low frequency supply noise and use of both theinverting input 1421 and non-inverting input 1422 of opamp 1420. In thismanner, opamp 1420 noise impact is reduced by approximately 2× byleveraging both the inverting and non-inverting gain paths.

The DC gain of the inverting path of the opamp corresponds to the ratioof the resistor across the feedback to the input resistor−(r13/(r10+r11)), while the noninverting path has DC gain of(1+r13/(r10+r11)). In the case where the magnitude of the DC gain of theinverting path is significantly larger than 1, then the magnitude of theDC gain of the noninverting path will have similar magnitude. As such,any common-mode signals such as supply noise in high gain PD cells 1425and 1426 will be largely cancelled out. For example, if the DC gain ofinverting path has magnitude of 10, then the DC gain of the noninvertingpath has magnitude of 1+10=11. In this case, supply noise will beattenuated by approximately 90% assuming the supply noise has the sameeffect on both high gain PD cells 1425 and 1426. Thus, good supply noisecancellation is provided in a single ended system (as opposed to adifferential two output system) which is convenient for doing analogcontrol of a VCO since a VCO typically has a single ended control input.

This example provides the benefit of supply noise cancellation of lowfrequencies, and effectively gets more gain out of the opamp. If justthe inverting terminal is used, then gain is 10 (in this example),however, in this case there is the gain of −10 on the inverting inputand 11 on the noninverting, then the total gain of the lossy integratingpath is effectively doubled in comparison with value of 21. As such, theopamp output provides double the phase error signal compared to justusing either the inverting or noninverting path. This is importantbecause the noise from the opamp is gained up by the noninverting pathgain so that doubling the gain of the phase error signal relative to thenoninverting path leads to roughly 2× improvement in Signal-to-Noiseratio at the opamp output. In effect, the opamp noise impact is reducedby about a factor of two. Therefore, this example provides the benefitof cancelation of low frequency supply noise by the integrating path andthe benefit of reduced impact of opamp noise in the system.

In this example, each PCC cell 1410, 1425, 1426 is operated on a 1.1Vregulated voltage Vreg. In another example, a different supply voltagemay be used. Each PCC cell 1410, 1425, 1426 and associated filternetwork can be optimized independently.

In this example, integrating path 1402 is described as being “lossy”integrator. To avoid saturation problems, feedback capacitor C13 isshunted by a feedback resistance R13. The parallel combination of C13and R13 behave like a practical capacitor which dissipates power, unlikean ideal capacitor. For this reason, a practical integrator is referredto as a lossy integrator. In another example, the amount of losscontributed by R13 may be selected based on other parameters to controlsaturation.

FIG. 15 is a schematic of a simple XOR phase detector cell 1500. In thisexample, a reference signal 1520 and a divided feedback signal 1521 areconnected to inputs of XOR gate 1501. Feedback signal 1521 is likefeedback signal 121 (FIG. 1) for PLL 100 (FIG. 1). Inverting buffer 1504provides buffered phase detect signal Vpdb 1507. Inverting buffer 1505provides on opposite phase detect signal Vpd 1506.

FIG. 16 is a schematic diagram of an example XOR differential high gainphase detector 1600 that uses simple XOR PD cells of FIG. 15 in place ofPCC cells. This structure is useful when a high frequency referencesignal, such as Ref 1520, is available. A way to increase PD gain is todecrease the time range that it takes to achieve a given voltage errorsignal from the PD after filtering, which is the case when operatingfrequency of the PD is increased. A typical reference frequency is lessthan a few hundred MHz, however, in this example the referenceoscillator runs at 2.5 GHz, while the VCO runs at multi-GHz. Since thereference frequency is very high, then don't need a PCC configurationwhere the reference frequency and the feedback divided signal aremultiples of each other and can instead use a simple XOR type PD, suchas PD 1500 (FIG. 15). In this example, the PD gain does not need to beincreased; instead, a simple implementation is desired in order to allowrobust operation at very high frequency. Due to the avoidance of narrowoutput pulses during steady-state operation, XOR PD provides a verylinear behavior in a system where delta sigma modulation is included inthe feedback loop as long as the instantaneous phase error deviation isnot so large as to create very small pulses at the PD output.

In this example, a wide band feed-forward (FF) path 1601 includes a PDcell 1610 coupled to FF filter 1412. Lossy integrating path 1602includes an opamp 1420 with an inverting input 1421 coupled to receive afiltered output from PD cell 1425 and a non-inverting input 1422 coupledto receive a filtered output from PD cell 1426.

Lossy integrating path 1602 also includes a frequency detection path1424 in which switch 1411 is configured to couple inverting input 1421to ground through resistor Rfd_lo when a signal FDlo asserts in caseswhere the frequency of an output signal, such as Out signal 122 (FIG. 1)is too low and in which switch 1412 is configured to couple invertinginput 1421 to Vreg through resistor Rfd_hi when a signal FDhi asserts incases where the frequency of the output signal is too high.

FF filter 1412 combines the output 1507 of PD cell 1610 and integratingpath 1602 to produce a control signal Vctrl on output node 1614. Controlsignal Vctrl is used to control a variable frequency oscillator thatproduces Out signal 122. In another example, additional filtering may beprovided for control signal Vctrl before being output on node 1414.

In this example, PD cells 1410, 1425, and 1426 are the same as PD cell1500 (FIG. 15), or another known or later developed PD cell. However,notice that output signal Vpd 1506 is coupled to inverting input 1421 ofopamp 1420, while the opposite polarity output signal Vpdb 1507 iscoupled on non-inverting input 1422. This allows cancellation of lowfrequency supply noise and use of both the inverting input 1421 andnon-inverting input 1422 of opamp 1420. In this manner, opamp 1420 noiseimpact is reduced by approximately 2× by leveraging both the invertingand non-inverting gain paths.

The DC gain of the inverting path of the opamp corresponds to the ratioof the resistor across the feedback to the input resistor−(r13/(r10+r11)), while the noninverting path has DC gain of(1+r13/(r10+r11)). In the case where the magnitude of the DC gain of theinverting path is significantly larger than 1, then the magnitude of theDC gain of the noninverting path will have similar magnitude. As such,any common-mode signals such as supply noise in high gain PD cells 1425and 1426 will be largely cancelled out. For example, if the DC gain ofinverting path has magnitude of 10, then the DC gain of the noninvertingpath has magnitude of 1+10=11. In this case, supply noise will beattenuated by approximately 90% assuming the supply noise has the sameeffect on both high gain PD cells 1425 and 1426. Thus, good supply noisecancellation is provided in a single ended system (as opposed to adifferential two output system) which is convenient for doing analogcontrol of a VCO since a VCO typically has a single ended control input.

This example provides the benefit of supply noise cancellation of lowfrequencies, and effectively gets more gain out of the opamp. If justthe inverting terminal is used, then gain is 10 (in this example),however, in this case there is the gain of −10 on the inverting inputand 11 on the noninverting, then the total gain of the lossy integratingpath is effectively doubled in comparison with value of 21. As such, theopamp output provides double the phase error signal compared to justusing either the inverting or noninverting path. This is importantbecause the noise from the opamp is gained up by the noninverting pathgain so that doubling the gain of the phase error signal relative to thenoninverting path leads to roughly 2× improvement in Signal-to-Noiseratio at the opamp output. In effect, the opamp noise impact is reducedby about a factor of two. Therefore, this example provides the benefitof cancelation of low frequency supply noise by the integrating path andthe benefit of reduced impact of opamp noise in the system.

In this example, each PD cell 1610, 1625, 1626 is operated on a 1.1Vregulated voltage Vreg. In another example, a different supply voltagemay be used. Each PD cell 1610, 1625, 1626 and associated filter networkcan be optimized independently.

FIG. 17 is a schematic of an example phase detector pulse generation(PG) circuit 1701, frequency detector circuit 1702, and bang-bang phasedetector circuit 1703. Reference frequency signal 120 and dividedfeedback signal 121 are provided as inputs to these circuits. PG circuit1701 generates phase detector control signals Up 924, Dn 925, and Gate926 that are used in the PCC cells described in more detail hereinabove.Frequency detector 1702 generates the FDhi and FDlo signals describedhereinabove in more detail when the frequency of the oscillator outputis outside of a selected range in order to achieve an initial lock ofthe PLL. Bang-bang PD circuit 1703 is used for DTC calibration and willbe described in more detail hereinbelow.

FIG. 18 is a timing diagram illustrating operation of the PG portion1701 of the example circuit of FIG. 17. In this example, the frequencyof reference frequency signal 120 is 1.25 GHz. The frequency of dividedfeedback signal 121 is 625 MHz, such that Ref 120 has a frequency of 2×Div 121. In another example, a larger multiple could be used, and also ahigher reference frequency. The 2× difference in frequency between Ref120 and Div 121 provides the equivalent of a 2× gain in the PD/PCC cell.Alternatively, in another example the frequency of Div 121 could beconfigured to be a multiple of the frequency of Ref 120.

Up/Dn pulses 924, 925 change width in opposite manner as a function ofphase error. This relationship provides high linearity even in thepresence of mismatch between Up/Dn loop filter paths. This is incontrast to prior techniques in which either Up or Dn pulses changeswidth independently.

FIG. 19 is a plot of noise in dBc/Hz vs offset frequency (Hz)illustrating noise in an example PLL In this example, a PLL 100 (FIG. 1)is equipped with the PCC blocks and loop filter circuit 1400 (FIG. 14)using the timing circuits 1701, 1702 (FIG. 17). In this example, thereference frequency is 1.25 GHz and the feedback frequency is 625 MHz.Low frequency noise from the regulated supply voltage (Vreg) is wellsuppressed as indicated by plot line 1902. Overall noise is indicated byplot line 1901. Overall jitter integrated from 12 kHz to 20 MHz is 46.0fs (rms).

In descried examples, a method of operating a phase locked loop (PLL) isdescribed. A first phase error signal is generated for a difference inphase between a reference signal and a feedback signal with first phasedetector cell 1425 (FIG. 14) having a gain polarity. A second phaseerror signal is generated for a difference in phase between thereference signal and the feedback signal with a second phase detectorcell 1426 having an opposite gain polarity. The first phase error signaland the second phase error signal are amplified by opamp 1420 (FIG. 14)and combined the results to form an integrated phase error signal. Athird phase error signal is generated for a difference in phase betweenthe reference signal and the feedback signal with a third phase detectorcell 1410 (FIG. 14) to form a feed-forward phase error signal. Thefeed-forward phase error signal is combined with the integrated phaseerror signal to form a control signal Vctrl 1414 (FIG. 14).

In described examples, a voltage-controlled oscillator (VCO) 108(FIG. 1) is operated responsive to the control signal Vctrl to generatean output signal Out 122 (FIG. 1). The frequency of the output signal iscontinuously monitored to determine if it is outside a target frequencyrange. A magnitude (value and/or sign) of the integrated phase errorsignal Vctrl is adjusted when the frequency of the output signal isoutside the target frequency range.

In described examples, the output of a divider 110 (FIG. 1) coupled tothe VCO is modulated with a delta sigma modulator. The feedback signalis delayed a varying amount responsive to the modulated output of thedivider by DTC 112 (FIG. 1).

In described examples, a first phase error signal is generated byapplying a first voltage to a first resistor-capacitor Rup, Cup1 (FIG.9A) for an amount of time proportional to a first phase difference toform a first RC node voltage, wherein the magnitude of the voltage isaugmented by a first charge pump 926 (FIG. 9A). A second voltage isapplied to a second resistor-capacitor Rdn, Cdn1 (FIG. 9A) for an amountof time proportional to a second phase difference to form second RC nodevoltage, wherein the magnitude of the voltage is augmented by a secondcharge pump 927 (FIG. 9A). The first RC node voltage and the second RCnode voltage are combined to form a combined RC node voltage by closingswitch 932 (FIG. 9A). The combined RC node voltage is transferred to afilter by switches 933, 934 (FIG. 9A). In some examples, the transfer ofthe combined RC node voltage to the filter is delayed for a period oftime by gate signals 13260, 13261 (FIG. 13B) to allow the combined RCnode voltage to stabilize.

Low Bandwidth, High Gain Phase Detector Examples

In the following examples, a differential switched RC front end is usedto cancel low frequency noise from a voltage regulator. A differentialfront end is combined with partial and fully differential loop filterand ADC (analog to digital converter). Gain of the loop filter is sethigh enough such that ADC noise impact is sufficiently reduced.

In some examples, a linear PD is augmented with a bang-bang detector andfrequency detector for reasonable lock-in time.

In some examples, a digital Delta-Sigma modulator is augmented to reducequantization noise at low frequencies without substantially increasingnoise folding by avoiding significant increase of quantization noisespectral magnitude at high frequencies.

FIG. 20 is a block diagram of an example voltage supply for an examplePLL. In a typical PLL system, a supply voltage 2001 is provided by acircuit, such as a bandgap circuit, that creates an accurate referencevoltage Vref. While Vref provides an accurate voltage value that isreasonably consistent across PVT variations, it is often prone to beingaccompanied by high noise and also does not provide sufficient outputcurrent to function as the supply for various circuits within theintegrated circuit including the PLL. As such, a voltage supplyregulator Vreg 2003 is utilized to provide sufficient output current forthe PLL and other blocks, and Vref 2001 is utilized as a referencevoltage for Vreg in order to achieve an accurate voltage across PVT. Thenoise present in Vref is filtered 2002 before being supplied to Vreg. Assuch, a typical supply regulator for the PLL has an output noisespectral density noise that is highest at low frequencies due to theimpact of band gap noise which is lowpass filtered.

FIG. 21 is an example plot of noise spectral density (V/rHz) vsfrequency (Hz) for the voltage supply regulator of FIG. 20, illustratingthe impact of Vref noise at lower frequencies.

FIG. 22 is a block diagram of an example feedback loop 2200 thatprovides a digital frequency ratio signal “OutN” on output node that isderived from comparison in phase/frequency of a high frequency bulkacoustic wave (BAW) oscillator 2201 and reference frequency Ftcxo. Inparticular, output signal OutN on node 2215 is the estimatedinstantaneous ratio of the frequency of BAW output signal 2202 and thefrequency of reference signal 2204, which in this example is provided bya temperature-controlled crystal oscillator (TCXO) 2203. BAW resonatorsfeaturing high operating frequency up to a few GHz and small size havebeen used for mobile applications such as filters in the RF front-end ofwireless transceivers for many years. The BAW resonator is apiezoelectric thin film resonator, which operates similarly to a quartzcrystal, is utilized by a BAW oscillator circuit to create a periodicoscillation signal. In this example, BAW oscillator 2204 operates at 2.5GHz.

Multi-modulus divider (MMD) 2206 divides BAW frequency signal 2202 byratio number N of feedback signal 2217 provided by digital delta sigmamodulator 2216. Div_early and div_late pulses are generated by MMD 2206,as illustrated in FIG. 23. In this example, delta sigma modulator 2216is clocked by div_late pulse 2219.

Phase detector 2208 uses reference signal 2204 and div_early anddiv_late pulses to generate phase difference signals including up pulse2209 and down pulse 2210 in response to the timing relationship betweenreference signal 2204 and the div_early and div_late pulses.

Phase to digital converter (P2DC) 2212 produces a digital output value2213 responsive to up pulse 2209 and down pulse 2210. Digital loopfilter 2214 filters digital value 2213 to produce output signal OutN onnode 2215.

In this example, initial lock-in time is improved by a “bang-bang” (BB)loop 2220, 2221 that will be described in more detail hereinbelow. TheBB loop augments the system with an extra phase detector when it isinitially settling. The BB loop provides an error signal to drive thesystem. Once the system locks, the BB loop drops out in activity anddoes not affect noise, etc.

In this example, delta sigma 2216 is designed to reduce delta sigmanoise impact without aggravating noise folding, as will be described inmore detail hereinbelow.

FIG. 23 is a timing diagram illustrating timing signals generated by MMD2206 (FIG. 22) and PD 2208 (FIG. 22).

FIG. 24 is a schematic of an example P2DC 2412 for a low BW feedbackloop, such as feedback loop 2200 (FIG. 22A).

Module 2412 includes switched resistor phase to charge converters (PCC)2425, 2426 that are configured in a differential manner. Each PCC 2425,2426 includes two switches, such as switches 2451, 2452 that arecontrolled by Up pulse signal 2209 and Dn pulse signal 2210,respectively. In this example, switches 2451, 2452 are each implementedas an FET transistor.

Differential loop filter 2401 includes opamp 2420. An output from PCC2425 is coupled to inverting input 2421 of opamp 2420 and an output fromPCC 2426 is coupled to non-inverting input 2422 of opamp 2420. Noticethat signals Up 2209 and Dn 2210 received from PD 2208 (FIG. 22A) arereversed between PCC 2425 and 2426.

Anti-alias filter 2430 attenuates frequencies above the Nyquist samplingrate of analog to digital converter (ADC) 2431 to eliminate aliasing.

ADC 2431 converts the amplified output from opamp 2420 into a digitalvalue that is output on node 2215. Such a digital value is useful for adigital phase locked loop (DPLL).

In this example, the differential configuration suppresses low frequencynoise on the regulated supply voltage Vreg and reduces the impact ofnoise produced by opamp 2420, as described in more detail for opamp 1420(FIG. 14).

FIG. 25 is a simulation model for the resistor switching section of FIG.24, such as switching section 2426. A value for equivalent resistor Rdet2501 is given by expression (6), though this expression is approximatein that Rdet can be reduced by the impact of parasitic capacitance innetworks 2425 and 2426 in FIG. 24.

$\begin{matrix}{R_{\det} = \left. {\frac{2T_{ref}}{T_{span}}R_{up}}||R_{dn} \right.} & (6)\end{matrix}$

Block 2508, which corresponds to the DC gain from phase error to Verror,indicates that the DC gain of phase detector 2412 is increased by theratio of the period of reference signal 2204 (FIG. 22) and the time spanTspan between the rising edges of Div_early and Div_late.

The DC gain of the loop filter circuit that is fed by Verror changes asa function of reference (TCXO) frequency based on expression (7), whereRdet is given by expression (6). Lower frequency for reference signal2204 leads to increased Rdet and therefore lower DC gain. Higherfrequency for reference signal 2204 leads to reduced Rdet and thereforehigher DC gain.

DC gain of loop filter=1+2*R _(fb)/(R _(det) +R _(neg))   (7)

The input to ADC 2431 (FIG. 24) and the output of opamp 2420 each have alimited voltage range. This leads to a tradeoff between effective phaseerror resolution and effective phase error range. Effective phase errorrange must be wide enough to accommodate jitter (including Delta-Sigmadithering). Effective phase error range is influenced by the ADC opampvoltage range, PD gain, and loop filter gain (expression (7)).

FIG. 26 is a schematic of an example fully differential P2DC 2600 for alow BW feedback loop, such as feedback loop 2200 (FIG. 22). In thisexample, phase to charge converters (PCC) 2425, 2426 are configured in adifferential manner and coupled to two separate opamps 2420, 2620.

Differential loop filter 2601 includes opamps 2420 and 2620. An outputfrom PCC 2425 is coupled to inverting input 2421 of opamp 2420 and anoutput from PCC 2426 is coupled to non-inverting input 2422 of opamp2420. Similarly, an output from PCC 2425 is coupled to non-invertinginput 2622 of opamp 2620 and an output from PCC 2426 is coupled toinverting input 2621 of opamp 2620. Notice that signals Up 2209 and Dn2210 received from PD 2208 (FIG. 22A) are reversed between PCC 2425 and2426.

An output 2423 from opamp 2420 and an output 2623 from opamp 2620 arecoupled to inputs of differential ADC 2631. ADC 2631 quantifies thedifference in voltage appearing on signal lines 2423 and 2623 andconverts it to a digital output. The output of ADC 2631 is then providedon output node 2215. ADC 2631 may be fully differential orpseudo-differential.

FIG. 27 is a schematic of an example fully differential P2DC 2700 for alow BW feedback loop, such as feedback loop 2200 (FIG. 22). In thisexample, phase to charge converters (PCC) 2425, 2426 are configured in adifferential manner with differential loop filter 2701 that includes asingle differential opamp 2720. An output from PCC 2425 is coupled toinverting input 2721 of opamp 2720 and an output from PCC 2426 iscoupled to non-inverting input 2722 of opamp 2720. Opamp 2720 providesdifferential outputs 2723, 2724 that are coupled to differential ADC2631. ADC 2631 quantifies the difference in voltage appearing signallines 2723 and 2724 and converts it to a digital output. The output ofADC 2631 is then provided on output node 2215. ADC 2631 may be fullydifferential or pseudo-differential.

FIG. 28 is a schematic of an example alternative switched resistor phaseto charge converter. In this example, switched PCC 2825 that has asingle resistor Rdet1 can replace PCC 2425 that has two resistors Rup1and Rdn1. Similarly, switched PCC 2826 that has a single resistor Rdet0can replace PCC 2426 that has two resistors Rup0 and Rdn0. Thisalternative configuration can be used in any of the previously describedsystems 2412, 2600, or 2700.

FIG. 29 is a schematic of an example alternative switch scheme for PCC2426, see FIG. 24. A similar configuration can be used in PCC 2425 (FIG.24). In this example, a buffer 2951 is inserted between switching FET2451 and Vreg and tracks the Up signal 2209. In this configuration thesupply voltage provided to switching transistor 2451 is provided by theoutput of buffer 2951. Therefore, when Up signal 2209 is inactive, avoltage that is approximately at ground potential is provided to switch2451. Similarly, a buffer 2952 is inserted between switching FET 2452and ground and inverts the Dn signal 2210. In this configuration thesupply voltage provided to switching transistor 2452 is provided by theoutput of buffer 2952. Therefore, when Dn signal 2952 is inactive, avoltage that is approximately Vreg potential is provided to switch 2452.In this manner, the off-resistance of switching circuit 2951, 2952 isincreased significantly. The on-resistance is increased only slightlydue to the resistance of buffers 2951, 2952.

FIG. 30 is a schematic illustrating example configurability options foran example PCC 3012 that has the same overall schematic as example PCC2412 (FIG. 24). It is beneficial to keep loop filter gain high enough sothat ADC 2413 quantization noise is well scrambled (i.e., so that atleast several ADC codes are exercised by noise or other signals).Referring to expressions (6) and (7), loop filter gain varies with theperiod of the reference frequency, Tref. Therefore, it is beneficial tomaintain sufficient loop filter gain as Tref varies by adjusting ortrimming various resistor and capacitor values in PCC 3012 asappropriate.

In this example, Rup0, Rup1, Rdn0, Rdn1, Rop_p, Rneg, Rfb, Cdet0, Cdet1and Cfb can each be individually adjusted using trimming switches andadditional resistors and capacitors in appropriate configurations, suchas connecting trimming components in series or in parallel. In thisexample, the trimming switches are controlled by a configurationregister (not shown) that is set by a control processor (not shown) forthe system. In another example, trimming may be controlled using knownor later developed techniques, such as: fusible links, erasableprogrammable read only memory (EPROM) bits, etc.

FIG. 31 is a block diagram and FIG. 32 is a timing diagram of an examplecircuit 3100 to generate early/late pulses 2218, 2219 in multi-modulusdivider 2206 as illustrated in FIG. 22. Multistage divider topology 3102provides a Div_early output pulse signal 2218 that is retimed from theDiv_In feedback signal 2217 (see FIG. 22). In this example, configurableshift register 3104 is utilized to accurately delay the Div_late outputpulse signal 2219 by a selected number of Div_in 2217 pulses. In thismanner, the length of time, Tspan, between a rising edge of Div_early2218 and a rising edge of Div_late 2219 is accurately set. In someexamples, shift register 3104 may be configured to allow Tspan to beadjusted by half cycles of Div_In feedback signal 2217. Known or laterdeveloped techniques can be used with multiplexors and registers tocontrol the configuration of delay register 3104 and thereby select avalue for Tspan.

FIG. 33 is a schematic and FIG. 34 is a timing diagram for an examplelinear phase detector 3300 that is included within PD 2208, see FIG. 22.In this example, flip-flop 3302 receives the Div_early 2218 pulse signalon a clock input and Div_late 2219 pulse signal on a reset input. Asolid “one” logic level is applied to a D input. Flip-flop 3302generates pd_pulse signal 3303 that is coupled to inputs on gates 3304,3306. Reference signal 2204 is coupled to a second input of gate 3304and an inverted version of reference signal 2204 is coupled to a secondinput of gate 3306. And-gate 3304 generates Dn pulse signal 2210, whileand-gate 3306 generates Up pulse signal 2209. In this example, anoptional delay module 3308 is included to delay Dn pulse 2210 by a smallamount so that the Up pulse and Dn pulse do not overlap. In thisexample, Tdelay is implemented using inverters. In another example,other types of known or later developed techniques or circuit elementsmay be used to produce a delay. In some examples, delay 3308 may beomitted if overlapping Up/Dn pulses are acceptable.

FIG. 35 is a schematic and FIG. 36 is a timing diagram for an examplelinear phase detector 3500 that is included within PD 2208, see FIG. 22.In this example, flip-flop 3502 receives reference signal 2204 on aclock input, a solid one logic level on a D input, and div_early on areset input. Flip-flop 3502 generates do pulse signal 2210. Flip-flop3504 receives reference signal 2204 on a clock input, a solid one logiclevel on a D input, and div_late on a reset input. Flip-flop 3504generates up pulse signal 2209. In this example, an optional delaymodule 3508 is included to delay Dn pulse 2210 by a small amount so thatthe Up pulse and Dn pulse do not overlap. In this example, Tdelay isimplemented using inverters. In another example, other types of known orlater developed techniques or circuit elements may be used to produce adelay. In some examples, delay 3308 may be omitted if overlapping Up/Dnpulses are acceptable.

FIGS. 37A-37E are timing diagrams illustrating example bang-bang (BB)timing signals generated by timing circuitry inside PD 2208 (FIG. 22)along with linear phase detector signals Up and Dn. As long as therising edge of reference signal 2204 falls within the Tspan window 3701,which is defined by the time between a rising edge of Div_early pulse2218 and a following rising edge of Div_late 2219, BB late 2223 and BBearly 2222 are inactive, as illustrated in FIGS. 37B, 37C, and 37D. FIG.37A illustrates an example case in which a rising edge of referencesignal 2204 occurs before the Tspan window 3701. In this case, BB earlysignal 2222 is activated. FIG. 37E illustrates an example case in whicha rising edge of reference signal 2204 occurs after the Tspan window3701. In this case, BB late signal 2223 is activated. Thus, in thisexample as long as both Up signal 2209 and Dn signal 2201 are activeindicating that the feedback loop 2200 (FIG. 22) is in lock, BB earlyand BB late are inactive. Note there are cases, such as encountered withthe PD circuit of 35, that Up and Dn could have activity outside of theTspan window without disturbing the relationship of BB early and BB latebecoming inactive outside of the Tspan window.

FIG. 38 is a schematic of an example circuit included within PD 2208(FIG. 22) to generate bang-bang signals. As long as the rising edge ofreference signal 2204 falls within the Tspan window 3701 (FIG. 37A),which is defined by the time between a rising edge of Div_early pulse2218 and a following rising edge of Div_late 2219, BB late 2223 and BBearly 2222 are inactive.

Re-timing flip-flops 3801, 3802 synchronize the timing of BB_early andBB_late to the div_late clock signal 2218, assuming sigma delta moduleis clocked by the div_late clock signal 2218. Re-timing flip-flops 3803,3804 synchronize the timing of BB_early and BB_late to the ADC clocksignal 2226, assuming ADC 2431 (FIG. 24) or ADC 2631 (FIG. 26 or 27)module is clocked by the ADC clock signal 2226.

Enhanced Digital Delta Sigma Modulator

As described hereinabove for FIGS. 1-5, noise folding may occur if DTC112 (see FIG. 1) is not included in PLL 100. Without DTC 112,nonlinearity in phase detector 102 leads to noise folding of delta sigmanoise 223, as indicated at 523 (FIG. 5). Such noise folding is avoidedby the use of DTC 112 which reduces the impact of dithering by deltasigma modulator 114 (see FIG. 1) on phase error. DTC 112 reduces phasevariation into phase detector 102 (FIG. 1) and also allows widebandwidth operation. However, a DTC adds complexity, power consumptionand area on an integrated circuit.

Delta-sigma (ΔΣ; or sigma-delta, ΔΣ) modulation is a method for encodinganalog signals into digital signals as found in an analog-to-digitalconverter (ADC). It is also used to convert high bit-count digitalsignals with relatively low frequency content into lower bit-count,higher-frequency digital signals in which the relatively low frequencycontent is preserved. For example, conversion of digital signals intoanalog as part of a digital-to-analog converter (DAC) as well asfractional-N frequency synthesizers may utilize Delta-Sigma modulation.The delta-sigma modulation technique is known, see for example:“Delta-sigma modulation,” Wikipedia, 9 August 2021 or later.

FIG. 39 is a block diagram of a 2nd order MASH digital delta-sigmamodulator 3901. The multi-stage noise shaping (MASH) digital structurehas a noise shaping property and is commonly used in digital audio andfractional-N frequency synthesizers. It includes two or more cascadedoverflowing accumulators, each of which is equivalent to a first-ordersigma-delta modulator. The carry outputs are combined through summationsand delays to produce a binary output, the width of which depends on thenumber of stages (order) of the MASH.

RC charging of a high gain PCC, such as PCC 2212 (FIG. 22), hasnonlinearity that causes noise folding of delta-sigma noise produced bydelta-sigma module 2216 (FIG. 22). 2^(nd) order MASH delta-sigma 3901often yields acceptably low noise folding, but not sufficient noiseshaping. However, a 3^(rd) order delta-sigma often yields unacceptablyhigh noise folding.

FIG. 40 is a block diagram of an example enhanced 2^(nd) order MASHdelta-sigma modulator 4001. In this example, 2^(nd) order MASHdelta-sigma 3901 is enhanced with a feedback loop that uses designparameters “K” and “a.” In this example, feedback block 4009 acts as adigital lowpass filter that extracts the low frequency quantizationnoise so that it can suppressed through the action of feedback. Theresulting feedback leads to a change in the DC gain of the closed loopsystem which is compensated by a cascaded gain block 4007.

FIG. 41 is an example noise model of the example enhanced delta-sigmamodulator 4001 of FIG. 40. In this example, expression (8) represents atransfer function of delta-sigma quantization noise 4010 to output node4005. An overall delta-sigma noise spectrum is represented by expression(9), where “n” is the MASH order of the delta-sigma. A signal transferfunction from input 4003 to output 4005 is represented by expression(10).

$\begin{matrix}{{H_{q}(z)} = \frac{1 - {az}^{- 1}}{1 + {K\left( {1 - a} \right)} - {az}^{- 1}}} & (8) \\{{S_{q}(z)} = {{1/1}2^{\star}{{H_{q}(z)}}^{2}{{1 - z^{- 1}}}^{2n}}} & (9) \\{{H_{sig}(z)} = \frac{\left( {1 + K} \right)\left( {1 - {az}^{- 1}} \right)}{1 + {K\left( {1 - a} \right)} - {az}^{- 1}}} & (10)\end{matrix}$

FIG. 42 is a plot illustrating simulation results in dB/Hz vs frequency(MHz) for the enhanced delta-sigma of FIG. 40 compared to conventionalorder 2 and order 3 MASH structures. Plot line 4201 represents 2nd orderMASH delta-sigma 3901 (FIG. 39). Plot line 4202 represents enhanced2^(nd) order MASH delta-sigma 4001 (FIG. 40). Plot line 4203 representsa 3^(rd) order MASH delta-sigma (not shown).

In this example, K is selected to be 3 and is set to achieve lowpassbandwidth in feedback loop 4009 (FIG. 40) of approximately 1/100 of theclock frequency. In this example, the high frequency noise stays aboutthe same, but an improvement of approximately 9 dB is observed at lowerfrequencies.

Phase Locked Loop Examples

FIG. 43 is a block diagram of an example frequency generating system4300 that includes high bandwidth analog phase locked loop 4301controlled by a low bandwidth feedback loop 2200 of FIG. 22. High BW PLL4301 is similar to high BW PLL 100 (FIG. 1). In this example, high BWPLL 4301 is locked to reference frequency Fbaw 2202 provided by BAWoscillator 2201 that provides a high frequency and low jitter. In thisexample, divider 4302 divides high frequency reference signal 2202 by afactor of four for simplicity, but high gain PD techniques discussedcould be applied and therefore lead to changes in the best choice ofthis divide value. In another example, a reference frequency may beprovided by another known or later developed technique, such as acrystal-based reference oscillator.

In this example, low BW feedback loop 2200 is also locked to Fbawreference frequency signal 2202 and to Ftcxo reference frequency signal2204 provided by a temperature-controlled crystal oscillator. In anotherexample, a reference frequency may be provided by another known or laterdeveloped technique, such as a crystal-based reference oscillator.

In this example, high BW PLL 4301 may include a high gain phase detector102 as described hereinabove in more detail. In this example, low BWfeedback loop 2200 may include a high gain PD 2208 as describedhereinabove in more detail.

In this example, digital processing logic 4310 receives OutN signal 2215from feedback loop 2200. OutN signal 2215 provides the value of theratio between the frequency of Fbaw reference signal 2202 and Ftcxoreference signal 2204. Processing logic 4310 converts this ratio into afraction value Nfrac 4311 that is provided to delta-sigma 114. By doingso, the ppm accuracy of Fvcol can be set according to Ftcxo, andsuppression of low frequency phase noise of the BAW can be achieved.

In this example, APLL 4301 is described. In another example, a digitalPLL may be used in place of APLL 4301.

FIG. 44 is a block diagram an example frequency generating system thatincludes the example frequency generating system 4300 of FIG. 43augmented by a digital PLL (DPLL) 4401. In this example, open loopcancellation of BAW low offset phase noise is provided by TCXO feedbackloop 2200 and analog PLL 4301, as described hereinabove in more detail.

In this example, DPLL 4401 provides closed loop tracking to Fref 4406 toprovide PPM accuracy and very low offset phase noise suppression. DPLL4401 includes time to digital converter (TDC) 4402, digital loop filter4403, multi-modulus divider 4404, and delta-sigma 4405.

In this example, APLL 4301 is described. In another example, a digitalPLL may be used in place of APLL 4301. Similarly, in this exampledigital PLL 4401 is described. In another example, an analog PLL may beused in place of digital PLL 4401.

Simulations

FIG. 45 is a plot of phase noise level (dBc/Hz) versus offset frequencyfor simulated operation of example noise model of FIG. 2 illustratingnoise folding effects of delta-sigma noise, see FIG. 5. In this example,plot line 4510 represents overall phase noise at output 122 (FIG. 1). Inthis example, the carrier frequency is 312.5 MHz, reference frequency120 (FIG. 2) is 40.0 MHz, divider 210 (FIG. 2) input is 2.5 GHz, BW is14.7 kHz. Significant degradation occurs at low frequencies due to noisefolding of delta-sigma quantization noise, but total noise remains belownoise targets for two example systems, as indicated at 4501, 4502.

FIG. 46 is a plot illustrating phase noise level (dBc/Hz) versus offsetfrequency for simulated operation of example system 4300 of FIG. 43. Inthis example, the carrier frequency is 312.5 MHz, BAW referencefrequency 2202 (FIG. 43) is 40.0 MHz, divider 110 (FIG. 43) input is 2.5GHz, BW is 14.7 kHz. In this example, plot line 4610 illustrates overallphase noise appearing on output 122 (FIG. 43). Plot line 4611illustrates delta-sigma noise with folding in TCXO loop 2200 (FIG. 43).Plot line 4612 illustrates BAW noise from a simulated parallel BAWoscillator 2201 (FIG. 43). Plot line 4613 illustrates quantization noisefor a 10-bit , 4.0 MHz ADC quantizer within TXCO loop 2200 (FIG. 43).Total noise remains below noise targets for two example systems, asindicated at 4501, 4502.

FIGS. 47A, 47B are plots illustrating operation of an example bang-bangcircuit generated by timing circuitry inside PD 2208 (FIG. 22). In thisexample, plot line 4701 illustrates a simulated step response of afractional ratio value 4311 (FIG. 43) feed into the delta-sigma 114(FIG. 43). Prior to the step, bang-bang output signals 2222, 2223 (FIG.22) are quiescent, as illustrated at 4702. After the step input, BBoutput signals 2222, 2223 are active for a small amount of time (ms) asindicated at 4703 in order to more quickly stabilize the ratio value.After a short period of time, the BB output signals 2222, 2223 again goquiescent, as indicated at 4704 once the ratio value has stabilized.

Other Embodiments

In described examples, high gain, high BW phase detectors and high gain,low BW phase detectors are presented. In described examples, these arecombined in various combinations to provide variable frequency systemsthat produce stable frequency signals that have low noise. In anotherexample, these components may be configured in various topologies toprovide enhanced low noise system performance.

In this description, the term “phase detector” is used to refer to acircuit that detects a difference in phase between a reference signaland a feedback signal. In some examples, a phase detector may include apulse generator timing circuit, such a PG circuit 1701 (FIG. 17). Inother examples, a phase detector may be a simple XOR gate as shown inFIG. 15. In some examples, a phase detector may include a “phase tocharge converter” (PCC) such as PCC 905 (FIG. 9). In some examples, aphase detector may include a phase to digital converter, such as phaseto digital converter 2212 (FIG. 22).

In described examples, an opamp is used in the PCC. In another example,another type of known or later developed amplifier configuration thathas an inverting and a non-inverting input may be used.

In this description, the term “couple” and derivatives thereof mean anindirect, direct, optical, and/or wireless electrical connection. Thus,if a first device couples to a second device, that connection may bethrough a direct electrical connection, through an indirect electricalconnection via other devices and connections, through an opticalelectrical connection, and/or through a wireless electrical connection.

Modifications are possible in the described embodiments, and otherembodiments are possible, within the scope of the claims.

What is claimed is:
 1. A feedback loop (FBL) comprising: phase detection(PD) circuitry having a reference input to receive a reference frequencysignal, a feedback input to receive a feedback signal, and phasedifference outputs; a phase to digital converter (P2DC) having inputscoupled to the phase difference outputs, and an output; the P2DCincluding: a first phase to charge converter (PCC) having a gainpolarity and having a first phase error output; a second PCC having anopposite gain polarity and having a second phase error output; adifferential loop filter having a filter output, the loop filter havingan amplifier with an inverting input coupled to the first phase erroroutput and a non-inverting input coupled to the second phase erroroutput, the amplifier having an output; an analog to digital converter(ADC), the ADC having an input coupled to the output of the differentialloop filter, the ADC having an output to provide the output of the P2DC;and a feedback path coupled to the output of the P2DC, with an output ofthe feedback path coupled to the feedback input of the PD.
 2. The FBL ofclaim 1, wherein the amplifier is a first opamp, further comprising asecond opamp, the second opamp having an inverting input coupled to thesecond phase error output and a non-inverting input coupled to the firstphase error output, the second opamp having an output; and wherein theADC is a differential ADC, the differential ADC having a first inputcoupled to the output of the first opamp and a second input coupled tothe output of the second opamp.
 3. The FBL of claim 1, wherein theamplifier output is a differential inverting and non-inverting output,and wherein the ADC has differential inputs coupled to the differentialoutputs of the amplifier.
 4. The FBL of claim 1, further comprising ananti-alias filter coupled between the input of the ADC and the output ofthe differential loop filter.
 5. The FBL of claim 1, wherein the phasedetection circuitry further comprises a bang-bang detector configured tobe active when a phase difference between the reference frequency signaland the feedback signal is outside of a defined range.
 6. The FBL ofclaim 1, wherein the first PCC includes a resistor-capacitor (RC) nodewith a first resistor coupled from the RC node to a first voltage sourceby a first switch, a second resistor coupled from the RC node to asecond voltage source by a second switch, and a capacitor coupled fromthe RC node to the second voltage source, wherein the first switch iscontrolled by first phase difference output of the PD and the secondswitch is controlled by a second phase difference output of the PD. 7.The FBL of claim 1, wherein the first PCC includes a resistor-capacitor(RC) node with the RC node coupled to a first voltage source by a firstswitch, the RC node coupled to a second voltage source by a secondswitch, and a resistor and capacitor in series coupled from the RC nodeto the second voltage source, wherein the first and second switches arecontrolled by respective phase difference outputs of the PD.
 8. The FBLof claim 6, wherein the first switch is a field effect transistor (FET),further comprising a first buffer having an input coupled to a firstphase difference output of the PD, wherein an output of the first buffercouples the first reference voltage to the FET; and wherein the secondswitch is an FET, further comprising a second buffer having an inputcoupled to a second phase difference output of the PD, wherein an outputof the second buffer couples the second reference voltage to the FET. 9.The FBL of claim 6, further including a first trimming resistor coupledto the first resistor, a second trimming resistor coupled to the secondresistor and a trimming capacitor coupled to the capacitor.
 10. The FBLof claim 1, further comprising a delta sigma modulator having an inputcoupled to the output of the P2DC, and an output coupled to the PDfeedback input.
 11. The FBL of claim 10, wherein the delta sigmamodulator is a 2^(nd) order multi-stage noise shaping (MASH) structure,further comprising a feedback loop filter coupled from the output of thedelta sigma modulator to the input of the delta sigma modulator.
 12. Adelta-sigma modulator comprising a 2^(nd) order multi-stage noiseshaping (MASH) structure having an input and an output; and a feedbackloop filter coupled from the output of the delta sigma modulator to theinput of the delta sigma modulator.
 13. The delta-sigma modulator ofclaim 12, further comprising a gain block coupled to the input of thedelta sigma modulator.
 14. A frequency generating system, comprising: afirst reference oscillator having a first reference signal output; asecond oscillator having a second reference signal output; a feedbackloop having a first input coupled to the first reference signal and asecond input coupled to the second reference signal, the feedback loophaving a ratio output, wherein the feedback loop is configured toprovide a signal on the ratio output representative of a ratio of thefirst reference frequency to the second reference frequency; and anphase locked loop (PLL) having a first input coupled to the firstreference signal and a second input coupled to the ratio output of thefeedback loop, the PLL having an output for a variable frequency signal.15. The frequency generating system of claim 14, wherein the feedbackloop includes: phase detection (PD) circuitry having a first referenceinput to receive the first reference frequency signal, a secondreference input to receive the second reference frequency signal, afeedback input to receive a feedback signal, and phase differenceoutputs; a phase to digital converter (P2DC) having inputs coupled tothe phase difference outputs, and an output; the P2DC including: a firstphase to charge converter (PCC) having a gain polarity and having afirst phase error output; a second PCC having an opposite gain polarityand having a second phase error output; a differential loop filterhaving a filter output, the loop filter having an amplifier with aninverting input coupled to the first phase error output and anon-inverting input coupled to the second phase error output, theamplifier having an output; an analog to digital converter (ADC), theADC having an input coupled to the output of the differential loopfilter, the ADC having an output coupled to the output of the P2DC; anda feedback path coupled to the output of the P2DC, with an output of thefeedback path coupled to the feedback input of the PD.
 16. The system ofclaim 15, wherein the amplifier is a first opamp, further comprising asecond opamp, the second opamp having an inverting input coupled to thesecond phase error output and a non-inverting input coupled to the firstphase error output, the second opamp having an output; and wherein theADC is a differential ADC, the differential ADC having a first inputcoupled to the output of the first opamp and a second input coupled tothe output of the second opamp.
 17. The system of claim 15, wherein theamplifier output is a differential inverting and non-inverting output,and wherein the ADC has differential inputs coupled to the differentialoutputs of the opamp.
 18. The system of claim 15, further comprising ananti-alias filter coupled between the input of the ADC and the output ofthe differential loop filter.
 19. The system of claim 15, wherein thephase detection circuitry further comprises a bang-bang detectorconfigured to be active when a phase difference between the referencefrequency signal and the feedback signal is outside of a defined range.20. The system of claim 14, wherein the PLL includes: a first phasedetector cell (PD) having a gain polarity, the first PD cell having areference input coupled to the first reference signal output, a feedbackinput coupled to the ratio output, and a phase error output; a second PDcell having an opposite gain polarity, the second PD cell having areference input coupled to the first reference signal output, a feedbackinput coupled to the ratio output, and a phase error output; and a loopfilter having a filter output, the loop filter having a feedforward pathand an integrating path coupled to the filter output; the feedforwardpath having a third PD cell, the third PD cell having a reference inputcoupled to the reference signal output, a feedback input coupled to theratio output, and a phase error output AC-coupled to the filter output;the integrating path including an opamp having an inverting inputcoupled to the first PD cell phase error output and a non-invertinginput coupled to the second PD cell phase error output, the opamp havingan output coupled to the filter output.